Electrical circuit tracing and identifying apparatus and method

ABSTRACT

A transmitter that is electrically connected to a circuit and an associated receiver is used to identify an element of that circuit in the presence of other circuits. This may be used in AC or DC systems that are powered or not to identify or trace circuit elements such as lines, junctions, switches, fuses, or breakers. One version that identifies the circuit breaker of the circuit connected to the transmitter in a powered AC distribution system.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates to electrical-test equipment usedfor tracing conductors and identifying electrical circuit elements.

[0003] 2. Discussion of Related Art

[0004] Electrical work often requires identifying elements of a circuitor tracing a circuit behind a wall or other obstruction. For example, anelectrician may wish to determine which circuit breaker is attached to aparticular wall outlet so that repairs may be made. By identifying theproper circuit, the electrician can de-energize just a single circuitbefore performing the repairs. Typically, the electrician prefers not toshut down equipment attached to other circuits. Alternatively, theelectrician may wish to trace a hidden wire along a wall to locate aconvenient place to add another outlet.

[0005] Some devices for locating and identifying electrical circuits usea transmitter and a receiver. A transmitter 10 induces a current signalon the circuit in question. A receiver 100 senses the induced signal.

[0006]FIG. 1 illustrates how a transmitter and receiver are used with apower distribution system. A transmitter 10 is physically connected to acircuit 11 in question. The transmitter 10 may be connected to thecircuit 11 by, for example, plugging the device into an outlet (asshown) or using jumper wires. The transmitter 10 induces an electricalcurrent signal in the circuit 11 in question. The circuit II radiateselectromagnetic radiation 20 along its path. The receiver 100 receivesthe electromagnetic radiation 20 emanating from the circuit 11 inquestion.

[0007] The receiver 100 may be used to identify a circuit breaker fuse13 connected to circuit 11 or may be used to trace hidden wires ofcircuit 11. To identify a circuit breaker, the electrician scans thereceiver 100 over a circuit breaker panel 12 containing multiple circuitbreaker fuses 13 and 14. Circuit breaker fuse 13 is directly connectedto circuit 11 while circuit breaker fuses 14 are connected to othercircuits 15. As the receiver 100 passes over circuit breaker fuse 13,the receiver 100 alerts the electrician. To trace a hidden wire, theelectrician passes the receiver 100 over the area suspected ofconcealing the circuit 11. The receiver 100 provides the electricianwith a signal strength indication of received electromagnetic radiation20.

[0008] Some devices for identifying and tracing electrical systems uselow-frequency, short duration signals. They use the line frequency of 50Hz or 60 Hz. The transmitter sends a short duration pulse that lasts forapproximately 10 microseconds. Due to the nature of the transmittedpulse, the frequency spectrum is very wide and an associated receiver isrequired to sense a wide-bandwidth radiated signal. For examples oflow-frequency, short duration pulse transmitters and wide-bandwidthreceivers, see U.S. Pat. No. 4,556,839, U.S. Pat. No. 4,906,938, U.S.Pat. No. 5,497,094, and U.S. Pat. No. 5,969,516, herein incorporated byreference.

[0009] Other devices for identifying and tracing electrical systemsmodulate a signal on a high-frequency carrier. Their carrier frequenciesrange from approximately 3950 Hz to approximately 200 kHz. Ahigh-frequency carrier has the advantage that the transmitter signaleasily couples to the receiver. For examples of transmitters andreceivers sending and sensing carrier signals modulated on ahigh-frequency, see U.S. Pat. No. 4,491,785, U.S. Pat. No. 4,642,556,U.S. Pat. No. 4,801,868, U.S. Pat. No. 5,418,447, U.S. Pat. No.5,422,564, and U.S. Pat. No. 6,163,144, herein incorporated byreference.

[0010] Known devices either: (1) use a manual calibration system thatrequires the electrician to adjust the sensitivity of the receiver; or(2) require the electrician to remember the strongest signal sensed as ascan is performed.

[0011] As described below, these known devices give false-positiveindications for several reasons. For example, (1) the signal from thetransmitter couples to adjacent circuits; (2) a load on another circuitmasquerades as the transmitted signal; and (3) the electrician fails toproperly calibrate the device.

[0012] A receiver 100 can give a false-positive indication when a signal20 from a transmitter 10 couples to adjacent or neighboring circuits 15.Electromagnetic radiation 20 radiates from the target circuit 11carrying the transmitted signal to neighboring circuits 15 thus inducingcurrent on the neighboring circuits. Coupling from the target circuit 11to neighboring circuits draws energy away from the target circuit 11.The magnitude of the signal coupled to a neighboring circuit 15 relatesto the transmitted signal's carrier frequency. The higher the carrierfrequency, the more easily the signal couples to other circuits. Achange to the carrier frequency causes a proportional change to themagnitude of the coupled signal. The non-target neighboring circuits 15re-radiate the coupled modulated signal and thus may lead tofalse-positive indications.

[0013] A receiver 100 can give a false-positive indication when a loadon another circuit 15 masquerades as the transmitted signal. Loads onother circuits 15 might generate noise that may be miss-interpreted as asignal from the transmitter 10. For example, power modulating devices,such as switching power supplies, light dimmers, and motor controllers,generate noise that a receiver 100 might erroneously identify as asignal from the transmitter 10. Some power modulating devices referencethe power line voltage and frequency when generating power.Consequently, these devices may create extraneous current noise atmultiples or harmonics of the power line frequency. A receiver 100 mightnot be immune to this current noise from active loads and mayerroneously determine that this noise is a signal sent by thetransmitter 10.

[0014] To address the noise immunity problem described above, sometransmitters use a modulation scheme that the receiver automaticallyrecognizes. Some devices modulate a low-frequency signal on ahigh-frequency carrier. These devices rely on the electrician toperceive the difference between a transmitter's signal and noisegenerated by loads. These devices offer visual and audio indicators thatpulse at the low-frequency signal rate. See, for example, U.S. Pat. No.4,642,556, U.S. Pat. No. 5,418,447, U.S. Pat. No. 5,422,564 and U.S.Pat. No. 6,163,144, herein incorporated by reference. High-frequencycarriers used by these systems more often exhibit detectableintercircuit coupling. By the selection of a high-frequency carrier,these systems inherently fail to address the problem of a transmittedsignal on a target circuit 11 coupling to adjacent circuits 15.

[0015] A receiver can also give a false-positive indication when theelectrician fails to properly calibrate the device. To calibrate somedevices, the electrician manually adjusts the gain of the receivedsignal using a sensitivity adjustment. By reducing the sensitivity,fewer signals are detected. The electrician take readings from each ofthe candidate elements while continually adjusting the calibrationcontrol until only one indication is obtained. Similar devices take adifferent approach that includes a thermometer-type visual display andvariable-volume audio indicator. The electrician is instructed toremember the largest signal observed and to take this signal as theidentified target circuit. See, for example, U.S. Pat. No. 6,163,144,herein incorporated by reference. These system rely on human experienceand skill to properly detect circuits.

[0016] Thus, there is a desire and need for a device and method capableof tracing conductors and identifying electrical circuit elements with areduced false-positive error rate.

SUMMARY

[0017] Embodiments of the present invention provide an improvedelectrical circuit tracing and identifying apparatus and method.Specifically, according to some embodiments of the present invention, amethod and apparatus provide fewer false-positive indications than dopresently known devices.

[0018] To reduce false-positive indications, some embodiments of thepresent invention transmit and receive a mid-range carrier frequencybetween 120 Hz and 3900 Hz. Using a mid-range carrier frequency reducescoupling to adjacent circuits.

[0019] To reduce false-positive indications, some embodiments of thepresent invention locate a carrier frequency between a pair of adjacentharmonics of the power line frequency. Locating a carrier frequencybetween harmonics of the power line frequency mitigates the confusionreceivers have in distinguishing between a transmitted signal andsignals generated by other loads.

[0020] To reduce false-positive indications, some embodiments of thepresent invention use a time-variant filter. The time-variant filterintegrates over an integral number of power line cycles to eliminateresponse at harmonics of the power line frequency to reduce confusionbetween the transmitted signal and signals generated by other loads.

[0021] To reduce false-positive indications and to reduce errors due toerroneous calibration by the electrician, some embodiments of thepresent invention automatically compare the varying levels of receivedsignals. By comparing received signal levels, a device automaticallycalibrates itself.

[0022] To reduce false-positive indications, some embodiments of thepresent invention implement a phase switching process. Phase switchinghelps to concentrate the spectral components of the transmitted signalabout the carrier frequency.

[0023] The present invention is better understood upon consideration ofthe detailed description below and the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0024]FIG. 1 illustrates how a transmitter and receiver are used with apower distribution system.

[0025]FIG. 2 is a block diagram of a transmitter in accordance with someembodiments of the present invention.

[0026]FIGS. 3A through 3D show a schematic diagram of aswitched-resistive load transmitter and voltage, conductance and currentwaveforms in accordance with the some embodiments of the presentinvention.

[0027]FIGS. 4A through 4E show a schematic diagram of anotherswitched-resistive transmitter and voltage, conductance and currentwaveforms in accordance with the some embodiments of the presentinvention.

[0028]FIG. 5 shows show a schematic diagram of a current sourcetransmitter in accordance with the some embodiments of the presentinvention.

[0029]FIGS. 6A through 6D show voltage, conductance and currentwaveforms in accordance with the schematic diagram of FIG. 5.

[0030]FIGS. 7A through 7B show duty-cycle conductance and currentwaveforms in accordance with the schematic diagram of FIG. 3A.

[0031]FIGS. 8A through 8E show phased-switched conductance and currentwaveforms in accordance with the schematic diagram of FIG. 3A.

[0032]FIGS. 9A through 9C show a schematic diagram of a DCcoupled-inductor transmitter and associated wave forms in accordancewith some embodiments of the present invention.

[0033]FIGS. 10A through 10E show a schematic diagram of an ACcoupled-inductor transmitter and associated wave forms in accordancewith some embodiments of the present invention.

[0034]FIGS. 11A through 11C show a schematic diagram of a current-pumpresonant-circuit transmitter and associated wave forms in accordancewith some embodiments of the present invention.

[0035]FIGS. 12A through 12C show a schematic diagram of a transmitterand associated wave forms in accordance with some embodiments of thepresent invention.

[0036]FIGS. 13A and 13B are schematic diagrams of a resonant-circuittransmitter in accordance with some embodiments of the presentinvention.

[0037]FIGS. 14A through 14D are block and schematic diagrams ofreceivers in accordance with some embodiments of the present invention.

[0038] In the present disclosure, like objects that appear in more thanone figure are provided with like reference numerals.

DETAILED DESCRIPTION

[0039]FIG. 1 illustrates how a transmitter 10 and a receiver 100 areused with a power distribution system. In some embodiments of thepresent invention, a transmitter 10 directly connects to a powerdistribution system through a wall outlet. In some embodiments, thetransmitter 10 directly connects to the distribution wiring by, forexample, jumper wires. The transmitter 10 then energizes a circuit 11 ofthe power distribution system. A receiver 100 detects electromagneticradiation 20 induced by the transmitter 10. The receiver 100 senses theelectromagnetic radiation 20 emanating from the energized wires of thecircuit.

[0040] With proper modifications, some embodiments of the presentinvention may be used to identify circuit elements or trace circuits onpowered or un-powered lines. Powered lines may carry either alternatingcurrent (AC) or direct current (DC) at a low or high voltage.

[0041] For un-powered lines, power is disconnected from the circuitbreaker box 12. The hot and neutral conductors are temporarily connectedtogether to create a closed circuit within the local distributionsystem. For the un-powered cases, the transmitter 10 requires aninternal power supply. For the powered cases, the transmitter 10 eithermay include an internal power supply, or may tap power directly from thecircuit 11 connected to the power distribution system.

[0042] Some aspects of the present invention are directed towards animproved transmitter used to induce a signal along trace wires and incircuit breakers. Some aspects of the present invention are directedtowards an improved receiver used to sense a transmitted signalemanating from wires and circuit breakers. Some circuits used in atransmitter include a dissipative load such as a resistive load or acurrent source. Some circuits include a reactive load such as a coupledinductor or an LC resonant circuit.

[0043]FIG. 2 is a block diagram of a transmitter in accordance with someembodiments of the present invention. The transmitter 10 includes anoscillator 31, an optional modulator 32, an optional phase inverter 33,and a signal generator 34. The transmitter 10 is electrically connectedacross one circuit 111 of a power distribution system.

[0044] The oscillator 31 of FIG. 2 provides a carrier signal with afrequency off, to the modulator. The carrier signal may be any function,for example, a sinusoidal wave or a square wave, with a mid-rangeprincipal frequency between 120 Hz and 3900 Hz and may be set between apair of adjacent harmonics of potential AC noise generators. By using amid-range frequency f_(c) below 3950 Hz, coupling among circuits issignificantly reduced. By using a mid-range frequency f_(c) betweenadjacent pairs adjacent of harmonics a 50 Hz, 60 Hz or 400 Hz powersystems, discrimination and detection by a receiver 100 is enhanced.Additionally, a transmitter 10 and a receiver 100 pair tuned betweenpairs of adjacent harmonics of multiple power systems may be used oneach of these types of power systems without retuning or adjustment. Forexample, a frequency f_(c) of 930 Hz lies relatively evenly spacedbetween pairs of adjacent harmonics of both 50 Hz and 60 Hz powersystems. Of course, many other frequencies that lie between pairs ofadjacent power line harmonics, such as approximately 570 Hz, 630 Hz, 870Hz, 1170 Hz, 1230 Hz and 1470 Hz, are also usable. The frequency f_(c)may be generated in a variety of ways well known in the art. Componentssuch as a crystal oscillator or a ceramic resonator may be used.Alternatively, the frequency may be synthesized from a powered linebeing sensed. Embodiments using a mid-range carrier frequency between120 Hz and 3900 Hz and lying between a pair of adjacent harmonics may bebetter understood with reference to FIGS. 3A through 6D below.

[0045] The modulator 32 of FIG. 2 is also optional. If the modulator isnot implemented, the carrier provided by the oscillator 31 passes to thephase inverter 33. If implemented, the modulator 32 performs additionalsignal conditioning, such as duty-cycle modulation. Duty-cyclemodulation periodically allows the oscillator signal to pass. As aresult, duty-cycle modulation can increase signal-to-noise ratio. Thepower dissipated by a transmitter 10 using duty-cycle modulation can bemade the same as a conventional transmitter, however, duty-cyclemodulation can produce a larger current signal. Duty-cycle modulationinduces a current signal for only a fraction of each transmitted cycle.The time for a receiver 100 to acquire and process the signal isincreased. Any number of other modulation schemes, for example, FSK,MSK, QPSK and spread spectrum, are possible. Duty-cycle modulation maybe better understood with reference to FIGS. 7A and 7B below.

[0046] The phase inverter 33 of FIG. 2 is incorporated into someembodiments of the present invention. The phase inverter 33 may beoperated before or after the modulator 32. The phase inverter 33performs phase inverting or phase switching. The instantaneous polarityof the line may be sensed and a phase inverse signal imposed on thetransmitted signal. The phase switching process may be implemented usingcombinational logic or an algorithm in a microcontroller ormicroprocessor. Phase switching prevents a carrier from being modulatedas a function of the line frequency and confers two benefits: first,more of the current goes into the target frequency rather than beingdivided into side-bands; and second, the primary signal frequencyreceived is not a function of line frequencies (e.g. 50, 60, or 400 Hz).Phase switching increases the first harmonics of the transmitted signal,thus increasing the probability of detection by the receiver 100 Phaseinverting or phase switching may be better understood with reference toFIGS. 8A through 8E below.

[0047] The signal generator 34 of FIG. 2 may be either a dissipativeload or a reactive load such as a switched-resistive load (as shown inFIGS. 3A and 4A), a switch-current source (as shown in FIG. 5), aswitched coupled inductor (as shown in FIGS. 9A and 10A), or an LCresonant circuit (as shown in FIG. 11A). The signal generator 34 of FIG.2 may be better understood with reference to the description below.

[0048] In the following equations, the relative phase between theswitched load g(t) and the line voltage v_(p)(t) is ignored formathematical convenience. The results focus on the relative magnitudesof various frequency components.

[0049]FIGS. 3A, 4A and 5A are schematic diagrams of transmitters inaccordance with some embodiments of the present invention. FIG. 3A showsa simple transmitter including a switched-resistive load transmitterwith conductance G. Conductance G is applied to the circuit by a switchS1 switching ON and OFF at a carrier frequency off, having a period ofT₀=1/f_(c). The carrier frequency f_(c) controls switch S1 such that theconductance G appears as represented in FIG. 3B. When the switch is notconducting (OFF), the conductance is zero. When the switch is conducting(ON), the conductance in G. A resistive load with conductance of G=(1/R)[mhos] is switched at frequency f_(c)=ω_(c)/2π. The resulting switchedconductance signal g(t)=G*square(ω_(c)) appears as a square wave with aconductance of G for the first-half of the duty-cycle and zero mhos forthe second-half of the duty-cycle.

[0050] Multiple factors are considered when selecting the value of thecarrier frequency f_(c). Carrier frequency f_(c) selected may beselected such that the value is lies between a pair of adjacentharmonics of various power systems. By selecting a frequency f_(c)between a pair of adjacent harmonics of a power system, the sensitivityrequirements for a receiver's detection circuitry are lessened. Byselecting a frequency f_(c) that lies between the various pairs ofadjacent harmonics of different power systems, the transmitter/receiverpair may be used in various geographic locations using different powerline frequencies.

[0051] Common power systems in the United States use a 60 Hz linefrequency. Harmonics of these systems lie at multiples of 60 Hz (e.g.,120 Hz, 180 Hz, 240 Hz, 300 Hz). Candidate carrier frequencies that liebetween a pair of adjacent harmonics of a U.S. system are approximately90 Hz, 150 Hz, 210 Hz, 270 Hz and so on. Common power systems in Europeuse a 50 Hz line frequency. Harmonics of these systems lie at multiplesof 50 Hz (e.g., 100 Hz, 150 Hz, 200 Hz, 250 Hz, 300 Hz). Candidatecarrier frequencies that lie between a pair of adjacent harmonics of aEuropean system are approximately 75 Hz, 125 Hz, 175 Hz, 225 Hz and soon. Candidate carrier frequencies that lie between a pair of adjacentharmonics of both 50 Hz and 60 Hz include approximately 80 Hz, 165 Hz,220 Hz, 270 Hz, 330 Hz, 380 Hz and so on.

[0052] Carrier frequency f_(c) selected may also be selected such thatthe value is a mid-range between frequency, that is, a frequency 120 Hzand 3900 Hz. The power spectrum from DC to 120 Hz of a typical powersystem often includes excessive noise. As the carrier frequency isincrease, a transmitted signal more easily couples to neighboringcircuits. Coupling to neighboring circuits is undesirable but somecoupling capability is necessary because a transmitted signal must atleast couple to a receiver's antenna in order for the receiver to sensethe transmitted signal. As the carrier frequency increases, thelikelihood that a transmitted signal will be detected also increases,however, at the cost of stronger coupling to neighboring circuits. Aboveapproximately 4000 Hz, inexpensive receivers can be built to receive thecoupling transmitted signal. Below approximately 3900 Hz, the typicaltransmitters and receivers are less efficient and are not sensitiveenough to detected the transmitted signal.

[0053] The more efficient transmitter/receiver pair of the presentinvention balances the need for coupling by way of improved transmissionand reception techniques. Preferably, a selected carrier frequency f_(c)lies between approximately 240 Hz and 2000 Hz and lies relatively evenlyspaced between adjacent harmonics of each 50 Hz, 60 Hz and 400 Hz powersystems.

[0054]FIG. 3C shows the voltage v_(p)(t) supplied by an AC power networkhaving a period of T_(p)=1/f_(p). Ignoring the relative phases, the ACvoltage signal v_(p)(t)=V_(p) cos(ω_(p)t) is a sinusoidal wave atfrequency f_(p)=ω_(p)/2π. In a typical 50 Hz power network, voltagev_(p)(t) has peaks at ±V_(p) equal to approximately to ±160 volts (110volts RMS) (e.g., Japan) or 1325 volts (220 volts RMS) (e.g., Europe).The period T_(p) of a 50 Hz signal is approximately 20 milliseconds(msec). In a typical 60 Hz power network, voltage v_(p)(t) has peaks at±V_(p) equal to approximately to ±160 volts. The period T_(p) of a 60 Hzsignal is approximately 167 msec.

[0055] Current i₂(t) through the resistive load creates a voltage dropof approximately V_(p) across the resistive load G. Applying Ohm's law,a current through the resistive load appears as a pulsed currenti_(p)(t) that follows the envelope of v_(p)(t) and is scaled by theproduct of voltage v_(p)(t) and switched conductance G through theswitch as shown in FIG. 4D. The current i_(p)(t) is induced on the pairof conductors making up the circuit 11 to be tested. The cumulativeresistance in the circuit 11 and of the non-idea components of thetransmitter 10 is represented by resistor R. The cumulative resistanceR, though not shown in subsequent schematic diagrams, is assumed to bepresent. The induced current i_(p)(t) creates electromagnetic radiation20 that a receiver 100 is designed to sense. Here, the analysis wasshown for the AC power case. A similar analysis may also be applied ifthe voltage source v_(p)(t) represents a DC power case or an un-poweredclosed circuit.

[0056]FIG. 4A shows an alternative embodiment to the simple transmitterof FIG. 3A. The circuit includes a switched-resistive load withconductance G that induces a current i₂(t) on to the line through afull-wave bridge rectifier having for diodes D. One advantage of using arectifier is that the bi-directional switch may be replaced with aunipolar switch S1 that can be implemented with a single FET, a singlepower MOSFET or a single bipolar transistor.

[0057]FIG. 4B shows the transmitter's conductance g(t) as the switchopens and closes. FIG. 4C shows the voltage v_(p)(t) supplied by thepower system. FIG. 4D shows the rectified voltage v₂(t) after thefull-wave rectifier and before the switched conductor. FIG. 4E shows thecurrent i₂(t) through the switch S1. Following Ohm's law, current i₂(t)is the product between the switched conductance g(t) and the rectifiedvoltage v₂(t). Current i₂(t) passes through the rectifier producingcurrent i_(p)(t), which induces electromagnetic waves 20 that emanatefrom circuit 11.

[0058]FIG. 4F shows the induced current i_(p)(t) for aswitched-resistive load transmitter applied to an AC power system. Thecurrent appears as a pulse train scaled by the envelope of the AC powerand by the conductance G of the resistive load.

[0059]FIG. 5 shows an alternative embodiment to the transmitter of FIG.4A. The resistor with conductance G in FIG. 4A is replaced with acurrent source 12. A switch S1 placed in series with the current source12 defines a switched-current source. A switched-current source may beimplemented with an emitter-follower style amplifier. The schematicdiagram of FIG. 5 is further described with reference to the timingdiagrams of FIGS. 6-8 described below.

[0060]FIG. 6A shows the current i₂(t) switching on and off at thecarrier frequency f_(c). FIGS. 6B and 6C show the voltages before andafter the rectifier as described above.

[0061]FIG. 6D shows the current i_(p)(t) that is induced on circuit 11by the transmitter 10. The current signal i₂(t) generated by the currentsource and switch passes through the rectifier and appears as a pulsetrain with each pulse having a constant magnitude but a sign equalingthe sign of the AC power system's voltage v_(p)(t). Unlike cases using aresistive load, the amplitude of current i_(p)(t) using a current sourceis a function of the sign and not of the amplitude of v_(p)(t).

[0062] According to some embodiments, for each of the circuits shown,the carrier frequency may be set at a mid-range and the carrierfrequency selected may be centered between a pair of adjacent harmonics.Additionally, duty-cycle modulation and phase-switching, shown below,may be used.

[0063]FIGS. 7A through 7B show duty-cycle conductance and currentwaveforms in accordance with the schematic diagram of FIG. 3A. Every T₂seconds, the modulator 32 of FIG. 2 switches on (enables) theconductance g(t) pulse train for a period of T_(ON) seconds, as shown inFIG. 7A. After T_(ON) seconds, the modulator 32 halts (disables) thepulse train for a period of T_(OFF) seconds. The ration of TON toT_(OFF) ranges from 1:1 to 1:10. A ration of T_(ON):T_(OFF)=1:1represents a 50% ON duty-cycle and allows a receiver to reduce theamount of time it needs to provide an update. A ration ofT_(ON):T_(OFF)=1:10 represents approximately a 10% ON duty-cycle andreduces power dissipation in a transmitter and allows the transmitter toplace a greater amount of power in the transmitted burst of pulses. Thesum of T_(ON) and T_(OFF) defines T₂. The period T₂ is also limited byestimated patience of an operator, that is, by the amount of time thatan operator will be expected to wait between updates. Additionally, theratio between TON and T₀ ranges from 50:1 to 500:1, thus allowing 50 to500 pulse of length T₀ in each ON period T_(ON).

[0064] For example, with a carrier frequency f_(c)=930 Hz, T₀ equalsapproximately 1 msec, TON ranges approximately from 50 msec to 500 msec,and TOFF ranges approximately from 50 msec to 1 second. In someembodiments, the carrier frequency f_(c) 930 Hz, TON equalsapproximately 250 msec and TOFF equals approximately 750 msec, thusresulting in a 25% duty-cycle.

[0065]FIG. 7B shows the resulting current i_(p)(t) induced on circuit11. If using switched conductance, the current i_(p)(t) will follow theenvelope of the voltage v_(p)(t) but scaled by the conductance G asshown. When the conductance g(t) is zero between groups of pulses, thecurrent i_(p)(t) will also be zero. The implementation of duty-cyclemodulation to the schematic of FIG. 3A is exemplary only and is notmeant to limit other implementations. If the conductance G of FIG. 3A isreplaced with the current source of FIG. 5, a similar current i_(p)(t)results; however, the signosoidal envelope is removed leaving a constantenvelope.

[0066]FIGS. 8A through 8E show phased-switched conductance and currentwaveforms in accordance with the schematic diagram of FIG. 3A. FIG. 8Ashows the voltage v_(p)(t) supplied by the power system. FIG. 8B showsma(t), which represents the sign of v_(p)(t). FIG. 8C shows thetransmitter's conductance g(t) before phase-switching. FIG. 8D shows thetransmitter's conductance g, (t) after phase-switching. Thetransmitter's conductance g_(φ)(t) is produced by the product ofm_(φ)(t) and g(t). FIG. 8E shows the current i_(p)(t) induced on circuit11. The current i₂(t) is the product between the switched conductanceg_(φ)(t) and the voltage v_(p)(t). Current i₂(t) passes through therectifier producing current i_(p)(t), which induces electromagneticwaves 20 in the circuit 11. If the current source of FIG. 5 replaces theconductor, the currents i₂(t) and i_(p)(t) do not follow the envelope ofv_(p)(t).

[0067] Without phased-switching, a classical modulator modulates asignal x(t) with a sinusoidal carrier signal s(t) at frequencyf_(s)=ω_(s)/2π. If the signal x(t) is also a sinusoidal wave but atfrequency f_(x)=ω_(x)/2π, where f_(x)<f_(s)/2, the resulting frequencydomain components of x(t)*s(t) have equal magnitude lying at frequencies(f_(s)±f_(x)) and −(f_(s)±f_(x)) If x(t) represents v_(p)(t), then, tofirst order, the square wave signal switched load g(t) at frequencyf_(c) equals the sinusoidal signal s(t) at frequency f_(s) whenfrequency f_(c)=f_(s). Therefore, without phased-switching, theclassical modulation produces frequency domain components at frequencies(f_(c)±f_(p)) and −(f_(c)±f_(p)).

[0068] Typical distribution system operate at a line at frequency f_(p)of 50 Hz, 60 Hz or 400 Hz. With a domestic or industrial power system inthe U.S., the line provides a signal with frequency with f_(p)=60 Hz. Aload frequency of f_(c)=870 Hz results in signals at (f_(c)±f_(p)) of810 Hz and 930 Hz. The energy in the resulting signal is equally splitand a transmitted frequencies are dependent on the line frequency f_(p)of the power system.

[0069] A Fourier series expansion of the conductance g(t) shown in FIG.3B is${g(t)} = {a_{0} + {\sum\limits_{n = 1}^{\infty}{a_{n} \times {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}$

[0070] where $\begin{matrix}{a_{0} = \frac{G}{2}} \\{a_{n} = \frac{2G}{\pi \left( {{2n} - 1} \right)}}\end{matrix}$

[0071] The current i,(t) induced on the line is: $\begin{matrix}{{i_{p}(t)} = {{v_{p}(t)} \cdot {g(t)}}} \\{= {\left( {V_{p} \cdot {\cos \left( {\omega_{p}t} \right)}} \right) \cdot \left( {a_{0} + {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}} \right)}} \\{= {{V_{p} \cdot a_{0} \cdot {\cos \left( {\omega_{p}t} \right)}} + {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot V_{p} \cdot {\cos \left( {\omega_{p}t} \right)} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}}\end{matrix}$

[0072] with${\sin \quad A\quad \cos \quad B} = {{\frac{1}{2}{\sin \left( {A + B} \right)}} + {\frac{1}{2}{\sin \left( {A - B} \right)}}}$

[0073] then $\begin{matrix}{{i_{p}(t)} = {{V_{p} \cdot a_{0} \cdot {\cos \left( {\omega_{p}t} \right)}} +}} \\{{\sum\limits_{n = 1}^{\infty}{\frac{a_{n} \cdot V_{p}}{2} \cdot \left\{ {{\sin\left( {{\left( {{2n} - 1} \right)\omega_{g}t} + {\omega_{p}t}} \right)} - {\sin\left( {{\left( {{2n} - 1} \right)\omega_{g}t} - {\omega_{p}t}} \right)}} \right\}}}}\end{matrix}$

[0074] The magnitude of the first harmonic is (a_(n)V_(p)/2) whichequals (GV_(p)/π). To reduce the impact of the line frequency, theswitching of the load can be modified by using phase inversion to shiftthe energy from frequencies at (f_(s)±f_(x)) to a frequency off.

[0075] By replacing conductance g(t) with phase-switched conductanceg_(φ)(t), the current i_(p)(t) induced on the line becomes:

g _(φ)(t)=m _(φ)(t)·g(t)

[0076] and $\begin{matrix}{{i_{p}(t)} = {{v_{p}(t)} \cdot {g_{\varphi}(t)}}} \\{= {\left( {V_{p} \cdot {\cos \left( {\omega_{p}t} \right)}} \right) \cdot \left( {{m_{\varphi}(t)} \cdot \left( {a_{0} + {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}} \right)} \right)}} \\{= {{V_{p} \cdot {m_{\varphi}(t)} \cdot a_{0} \cdot {\cos \left( {\omega_{p}t} \right)}} +}} \\{{{m_{\varphi}(t)} \cdot {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot V_{p} \cdot {\cos \left( {\omega_{p}t} \right)} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}}\end{matrix}$

[0077] with

m _(φ)(t)·cos(ω_(p)t)=|cos(ω_(p)t)|

[0078] then${i_{p}(t)} = {{V_{p} \cdot a_{0} \cdot {{\cos \left( {\omega_{p}t} \right)}}} + {{{\cos \left( {\omega_{p}t} \right)}} \cdot {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot V_{p} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}}$

[0079] with${V_{p} \cdot {{\cos \left( {\omega_{p}t} \right)}}} = {b_{0} + {\sum\limits_{n = 1}^{\infty}{b_{n} \cdot {\cos\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}$

[0080] then $\begin{matrix}{{i_{p}(t)} = {{a_{0} \cdot V_{p} \cdot {{\cos \left( {\omega_{p}t} \right)}}} +}} \\{{\left( {b_{0} + {\sum\limits_{n = 1}^{\infty}{b_{n} \cdot {\cos \left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}} \right) \cdot {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}}}} \\{= {{a_{0} \cdot V_{p} \cdot {{\cos \left( {\omega_{p}t} \right)}}} + {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot b_{0} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}} +}} \\{{\sum\limits_{n = 1}^{\infty}{b_{n} \cdot {\cos\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)} \cdot {\sum\limits_{m = 1}^{\infty}{a_{m} \cdot {\sin\left( {\left( {{2m} - 1} \right)\omega_{g}t} \right)}}}}}} \\{= {{a_{0} \cdot V_{p} \cdot {{\cos \left( {\omega_{p}t} \right)}}} + {\sum\limits_{n = 1}^{\infty}{a_{n} \cdot b_{0} \cdot {\sin\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}}} +}} \\{{\sum\limits_{n = 1}^{\infty}{\sum\limits_{m = 1}^{\infty}{a_{m}{b_{n} \cdot {\sin\left( {\left( {{2m} - 1} \right)\omega_{g}t} \right)} \cdot {{\cos\left( {\left( {{2n} - 1} \right)\omega_{g}t} \right)}\quad.}}}}}}\end{matrix}$

[0081] Therefore, the amplitude of the spectral component of i_(p)(t) atcog is (a₁b₀), which equals (2V_(p)/π)*(2V_(p)/π)=(4V_(p)G/π²). That is,when using phase switching the magnitude of the first harmonic is(4GV_(p)/π²). The ratio of the non-phase-switched first harmonicmagnitude (a_(n)V_(p)/2) and phase-switched first harmonic magnitude(4GV_(p)/π²) shows that phase-switching provides a 27% improvement inamplitude over an equivalent non-phase-switched implementation.

[0082] Similarly, for the current source circuit of FIG. 5, theamplitude of the first harmonics can be shown to be (2I_(o)/π) withoutphase shifting and (4I_(o)/π²) with phase shifting, thus providing atheoretical improvement of approximately 57%.

[0083] Features described above, namely: (1) use of a carrier frequencyset at a mid-range; (2) use of a carrier frequency centered between apair of adjacent harmonics; (3) use of duty-cycle modulation; (4) use ofphase-switching; may be applied in conjunction with a dissipative load(e.g., a resistive load or a current source, described above) or with areactive load (e.g., coupled inductors and LC resonant circuits,described below). These features may also be combined with one another.For example, a mid-range carrier frequency centered between a pair ofadjacent harmonics may use a circuit including a dissipative load or areactive load.

[0084] The techniques described above may be replaced or supplementedwith the two reactive techniques described below. The first reactivetechnique uses the magnetic field of mutually coupled inductors to storeand discharge energy and is shown for both the DC and AC cases. Thesecond reactive technique uses a series resonant circuit.

[0085]FIG. 9A is a schematic diagram of a DC coupled-inductortransmitter in accordance with some embodiments of the presentinvention. The voltage source v_(p)(t) is assumed to be a DC voltagesource. Two magnetically coupled inductors L1 and L2 are coupled withreverse polarities. A first chain includes an inductor L1 and a switchS1 connected in series. A second chain includes an inductor L2 and adiode D2 connected in series. The two chains are connected in parallelwith the DC power source having voltage v_(p)(t).

[0086]FIGS. 9B and 9C show switch position and current waveforms inaccordance with the schematic diagram of FIG. 9A. FIG. 9B shows theposition of switch S1 relative to FIG. 9C. The waveforms assume switchS1 is open (turned OFF) at time t<0, there is ideal coupling between L1and L2, and there is equality of the self-inductance. When S1 is closed(turned ON), a current i_(p)(t)=I₁, begins to flow with the indicatedpolarity. When S1 is subsequently opened (turned OFF), the collapsingmagnetic field causes a current i(t)=−I₂ to flow. From an energyperspective, energy accumulates in the magnetic field during the periodwhen the switch SI is closed (turned ON) and released when the switch S1is opened (turned OFF). A receiver 100 may be used to detectelectromagnetic radiation emanating from the circuit 11 as a result ofthe saw tooth current i_(p)(t). This coupled-inductor technique shownfor a DC case may be extended to an AC case source as show in FIG. 10A.

[0087]FIG. 10A is a schematic diagram of an AC coupled-inductortransmitter in accordance with some embodiments of the presentinvention. The voltage source v_(p)(t) is assumed to be an AC voltagesource. Two magnetically coupled inductors L1 and L2 are coupled withreverse polarities as with the DC case. A first chain includes aninductor L1, a switch S1 and a diode D1 connected in series. A secondchain includes an inductor L2, a switch S2 and a diode D2 connected inseries. The two chains are connected in parallel with the AC powersource having voltage v_(p)(t).

[0088]FIGS. 10B through 10E show switch position and current waveformsin accordance with the schematic diagram of FIG. 10A. FIG. 10B shows thevoltage v_(p)(t) supplied by an AC power network having a period ofT_(p)=1/f_(p). Again, the switching frequency (carrier frequency f_(c))is assumed to be substantially greater than the frequency of the powersignal. FIGS. 10C and 10D show the switching of switch S1 and S2. Whilethe polarity of v_(p)(t) is positive, switch S2 is held in the connectedposition and switch S1 is actively modulating the current by turning ONand OFF at the selected carrier frequency. While the polarity ofv_(p)(t) is positive, the circuit operates substantially as describedwith the DC case above. When the polarity of v_(p)(t) changes frompositive to negative, switch S1 is held in the connected position andswitch S2 begins actively modulating the current by turning ON and OFFat the selected carrier frequency. While the polarity of v_(p)(t) isnegative, the circuit operates equivalent via symmetry. FIG. 10E showsthe resulting current signal i_(p)(t) induced on the circuit 11.

[0089]FIG. 11A is a schematic diagram of a resonant-circuit transmitterin accordance with some embodiments of the present invention. Inaccordance with other embodiments of the present invention, a reactivesystem uses a series-resonant circuit to share energy storage betweenmagnetic and electrical fields. An inductor L is connected is series toa capacitor C. A current-source 12 is connected in series with a switchS1. The current-source 12 and switch S1 are connected in parallel acrossthe capacitor C. The power system provides an AC voltage v_(p)(t) andthe circuit induced a current i_(p)(t) on to the circuit 11. The currentsource 12 and switch S1 define a switched current source.

[0090] The carrier frequency f_(c) is selected as described above. Theinductor L and capacitor C of the LC resonant circuit are selected toresonate at the carrier frequency f_(c). The switched current source isswitched ON and OFF at the rate of the carrier frequency f_(c) and isused to inject a small amount of current into the LC circuit.

[0091]FIG. 11B shows the switch S1 ON and OFF transitions. Theduty-cycle of the switch S1 is selected such that the injected currentcompensates for the natural and parasitic resistive elements of thetransmitter 10 and circuit 11. If the switch is opened and closed at theresonant frequency f_(c) of the series LC circuit, the energy lost isreplenished. By adjusting the duty-cycle of the switch and the magnitudeof the current source, a current i_(p)(t) can be maintained at verynearly sinusoidal wave with frequency f_(c). In some embodiments theduty-cycle is approximately 3% ON.

[0092]FIG. 11C shows the current waveform in accordance with theschematic diagram of FIG. 11A. Generally, the LC circuit resonants atthe carrier frequency f_(c). In the ideal case, that is, withoutresistive losses, the LC circuit would continue to resonant oncestarted. Assuming the ideal case for the LC circuit, the currenti_(p)(t)=A sin(ω_(c)t), where f_(c)=2πω_(c) is the resonant frequency ofthe LC circuit and A is a scalar constant. The current i_(p)(t) in willremain A sin (ω_(c)t) for all time and there is no energy dissipated.

[0093] For the realistic case, wiring resistance, componentimperfections, parasitic resistances in the capacitor and inductor causesmall amounts of power to dissipate from the LC circuit. If leftunattended, the current envelope would slowly attenuate until no currentwas resonating in the LC circuit. If power is periodically injected intothe circuit with the switched current source, the decaying current isreplenished, thus maintaining a relatively constant current envelope.

[0094] The circuit shown in FIG. 11 works equally well with either DC orAC power sources. If the AC source frequency is much less than theresonant frequency. The only change is an addition to i of a current atthe AC source frequency. The fact that the signal current is notdependent on the value of the source voltage is beneficial for designsthat are used over a range of voltages.

[0095] The advantage of the series resonant circuit vis-à-vis thecoupled inductor circuit is that it is much cheaper. The disadvantage isthe limited opportunities for modulation—it functions best at a singlefrequency whereas the coupled inductor circuit can be modulated easily.Both share the same concept: alternate between energy-accumulation andenergy-sourcing to generate a current signal.

[0096]FIGS. 12A, 12B and 12C are block and schematic diagrams of atransmitter in accordance with FIG. 3C. The transmitter 10 of FIG. 12Aincludes a front-end filter 41, a phase extractor 42, a microcontroller43, a rectifier 44 and a switched load 45. The front-end filter 41filters high bandwidth noise. The phase extractor 42 tracks thefrequency and phase of the incoming AC voltage v_(p)(t). The extractedfrequency and phase information is passed to the microcontroller 43. Themicrocontroller 32 provides a signal to the switched load 45 to controlthe switching of the load's switch. If phase-switching is implemented,the microcontroller 43 can use the power frequency and phase informationto switch the phase of the switch control signal sent to the switchedload 45. The rectifier 44 provides a bi-polar current source that isswitched ON and OFF by the microcontroller 43.

[0097]FIGS. 12B and 12C show a detailed implementation of the currentsource transmitter of FIG. 5. A clip-clamp circuit which serves as phaseextractor 42 taps the input AC voltage v_(p)(t) and provides a squarewave with a high of +V_(pp) volts, a low of 0 volts and a frequency off_(p) to the microcontroller 43. The microcontroller 43 is programmed toprovide a switching signal that transitions at the carrier frequencyf_(c). If phase-switching is enabled, the microcontroller 43 adjusts theswitching signal before sending it to the switch. The switching signalpasses through resistor R12 to the switch. The current source 12 andswitch SI of FIG. 5 are replaced with the emitter-follower circuitshown. The switch, here implemented with transistor Q4, is controlledthrough the base of Q4 by the microcontroller's switching signal. Whenthe transistor Q4 is turned ON, current is drawn from the full-wavebridge rectifier 44 having for diodes D3 D4 D5 and D6. The current drawnappears as i_(p)(t) going into the transmitter 10. Additionally, the LEDD11 provides a visual indication to the electrician that the line ishot.

[0098]FIG. 12C is a further detailed version of FIG. 12B, and shows avoltage generator and a PIC processor. A shunt-voltage generator,comprised of diodes D1 and D2, resistor R1 and capacitor C1, may be usedas a reference for the clip-clamp circuit 42 and can be used to powerthe microcontroller 43. Also shown is a PIC12C508 microcontroller 43.The PIC12C508 processor is an 8-pin, 8-bit CMOS microcontrollermanufactured by Microchip Technology Inc. (2355 W. Chandler Blvd.,Chandler, Ariz. 85224). The processor synthesizes the carrier frequencyswitching signal, for example, at 930 Hz.

[0099]FIGS. 13A and 13B are schematic diagrams of a resonant-circuittransmitter in accordance with some embodiments of the presentinvention. The low pass filter of FIG. 13B has been replaced with aseries RLC circuit 50. The RCL circuit is designed to resonant at theresonant frequency f_(c). The switched current source 45 injects currentthrough the rectifier 44 into the RCL resonant circuit 50 to compensateenergy dissipated in the circuit. A resistor R11 may be used to keep aconstant current flowing through the circuit while the transistor Q4 isOFF.

[0100] In some embodiments, the microcontroller 43 synthesizes a 930 Hzcarrier switching signal from v_(p)(t). The emitter-follower is switchedat a rate of 930 Hz with an on-time about 3% of the period.

[0101]FIGS. 13A, 13B transmitter operates over a wide range of voltages.The lowest operating voltage is defined by both the diode drops and bythe low end of the emitter-follower amplifier's operating range. Thehigh operating voltage is defined by the maximum voltage rating of thecomponents. The components in FIG. 13B are designed for an AC voltagerange of 100 to 277 V_(rms) at 50 or 60 Hz.

[0102] The transmitters described above induce a current i_(p)(t) on thecircuit 11. A receiver 100 is required to sense the electromagneticradiation resulting from the induced current i_(p)(t). The receivertypically needs no physical contact with the circuit 11. The signal ismeasured by sensing the electromagnetic field 20 in circuit 11 with anunshielded inductor, hall-effect device, or flux gate. Passive- and/oractive-analog signal-conditioning circuits with a band-pass transferfunction centered at the signal carrier frequency f_(c) precede a signalisolator that produces a metric indicative of signal intensity.

[0103] In some embodiments, a two-step process generates the signalintensity metric. First, a raw metric is derived through for examplevery-narrow band-pass filtering, or asynchronous demodulating of theinput. Second, the raw metric is refined through integration for aperiod of time equal to an integral number of line frequency cycles toyield zero responses of line frequency harmonics. The integrator may beimplemented, for example, with an analog gated-integrator or withdigital arithmetic. Previous approaches to isolate the desired carrierfrequency used simple linear time-invariant filtering rather than thetime-variant technique disclosed here. These previous approaches do notprovide the high degree of immunity to interference as the presentinvention.

[0104] Additionally, a receiver 100 may include automatic calibration.In some embodiments, automatic calibration is performed by using acombination of memory and comparitors. The memory holds the value of thelargest measurement seen as the electrician initially scans thecandidates. Subsequently, the indicator(s) will show a positive resultat the target. The comparitor discriminates against the maximum andsub-maximum readings. The memory may be implemented, for example,digitally with conventional on-board or separate memory, or via analogcircuitry such as a sample-and-hold circuit. Similarly, the comparitormay be implemented with an analog comparator or with a digitalarithmetic processor.

[0105]FIGS. 14A through 14D are block and schematic diagrams ofreceivers in accordance with some embodiments of the present invention.A fluctuating magnetic field generated by the signal current produces apotential across the sensor 101 or transducer 201. The potential isconditioned then band-pass filtered with filter 102 or band pass filter202. A microcontroller 103, for example an PIC12C671, includes an ADC,which converts filtered result s(t) into a digital word. The magnitudeof the magnetic field intensity s(t) at carrier frequency f_(c) may beused as the raw metric. For example, at a carrier frequency f_(c)=930Hz, the raw metric may be computed as:${raw\_ metric} = {{{s(t)}} = \sqrt{\left( {{s(t)}{\sin\left( {2\quad {\pi (930)}t} \right)}} \right)^{2} + \left( {{s(t)}{\cos\left( {2\quad {\pi (930)}t} \right)}} \right)^{2}}}$

[0106] The raw metric may be integrated for a set period of time, forexample, 0.1 seconds, thus produce a refined metric that is zero at allfrequencies that are multiples of 10 Hz except 930 Hz. Averaging theresults from two integration operations then forms a final measurement 5times a second.

[0107] Each time a new measurement is performed it is compared to areference. If the measurement is less than a set percentage of thereference, then, if used, a red LED or other proper optical indicatorilluminates. If the measurement is greater than the set percentage ofthe reference, a green LED or other proper optical indicator illuminatesand/or a buzzer or other audio indicator sounds. The set percentage maybe for example between approximately 70 to 95%, such as 90%.Additionally, if the measurement is greater than the reference, then thereference is set to one-half of the sum of the current reference and thelast measurement. Alternatively, if the measurement is greater than thereference for two consecutive measurements, then the reference is set toone-half of the sum of the current reference and the last measurement.The process repeats, thus updating the reference value and providingvisual/audio indications on an ongoing basis as appropriate. The processdescribed implements automatic calibration in the receiver.

[0108] A method for finding an electrical circuit using a transmitterand receiver as described above includes connecting teh transmitter toan electrical circuit, inducing a modulated signal at a carrierfrequency onto the electrical circuit, sensing the electrical circuitwith a receiver which detects the carrier frequency, by generating ametric, saving the metric as a reference, generating a next metric,comparing the reference to the next metric, and if the next metric is aset percentage of the reference, generating an alert indication to anoperator, comparing the reference to the next metric; and if the nextmetric is greater than the reference, setting the reference to theaverage of the next metric and the reference; and repeating the acts ofgenerating the next metric, comparing and generating the alert, andcomparing and setting the reference.

[0109] Another method for finding an electrical circuit using atransmitter and receiver as described above includes providing a carrierfrequency between approximately 120 Hz and approximately 3900 Hz,modulating the carrier frequency, generating a signal across a modulatorand a load, coupling the signal to the electrical circuit; and detectingthe signal in the conductive circuit.

[0110] For identifying circuit elements such as junctions, breakers, andfuses in powered AC distribution systems all of the techniques apply andthe transmitter uses the available power to generate the signal byloading the line. The same methods may be used on a powered DC system(DC power systems are often used in places such as aircraft, ships,locomotives and associated rolling freight, road vehicles andspacecraft) but then only numbers 1 and 4 above confer any benefit.

[0111] For tracing a circuit in a powered AC system the automaticcalibration is deleted from the receiver or simply disabled in areceiver that can perform both identification and tracing. Then thesignal level is measured against an unchanging reference (supplied by afactory setting, manual calibration, or automatic calibration). Tracingis performed via a “closer/further” indication to the electrician byvisual and/or audio means. In this application all of the techniqueslisted are applicable except number 4. When tracing powered DC systemsthe same modifications apply and all of the techniques may be used butonly number 1 confers any benefit.

[0112] Both identification and tracing as described in the previous twoparagraphs can be extended to un-powered systems through the simpleexpedient of providing the transmitter with an appropriate power sourcein series with the signal generator of FIG. 2 and shorting the mainservice (see FIG. 1) of the distribution system in question.

[0113] The above detailed descriptions are provided to illustratespecific embodiments of the present invention and are not intended to belimiting. Numerous modifications and variations within the scope of thepresent invention are possible. The present invention is defined by theappended claims.

We claim:
 1. An electrical circuit finder for detecting a conductivecircuit comprising: a transmitter for transmitting an electrical signalto the conductive circuit and which includes: a modulator which operatesat a carrier frequency between approximately 120 Hz and approximately3900 Hz; and a load serially connected to the modulator; wherein theelectrical signal is generated across the modulator and the load and thetransmitter is electrically coupled to the conductive circuit; and areceiver which detects the electrical signal in the conductive circuit.2. The electrical circuit finder of claim 1, wherein the carrierfrequency is between an adjacent pair of harmonics of an AC powersystem.
 3. The electrical circuit finder of claim 2, wherein the carrierfrequency is between an adjacent pair of harmonics of 50, 60 or 400 Hz.4. The electrical circuit finder of claim 2, wherein the carrierfrequency is between approximately 240 Hz and approximately 2000 Hz. 5.The electrical circuit finder of claim 1, wherein the carrier frequencyis approximately 930 Hz.
 6. The electrical circuit finder of claim 2,wherein the modulator includes a non-linear circuit.
 7. The electricalcircuit finder of claim 6, wherein the non-linear circuit is a singleswitch.
 8. The electrical circuit finder of claim 6, wherein thenon-linear circuit includes a plurality of switches.
 9. The electricalcircuit finder of claim 2, wherein the modulator includes a linearcircuit.
 10. The electrical circuit finder of claim 9, wherein thelinear circuit is an amplifier.
 11. The electrical circuit finder ofclaim 10, wherein the amplifier includes an emitter-follower circuit.12. An electrical circuit finder for detecting a conductive circuit inan AC power system, the finder comprising: a transmitter fortransmitting an electrical signal to the conductive circuit and whichincludes: a modulator which operates at a carrier frequency, wherein thecarrier frequency is modulated by a sign function which represents asign of a voltage signal of the AC power system; and a load seriallyconnected to the modulator; wherein the electrical signal is generatedacross the modulator and the load; and a receiver which detects theelectrical signal in the conductive circuit.
 13. The electrical circuitfinder of claim 12, wherein the carrier frequency is betweenapproximately 120 Hz and approximately 3900 Hz; and wherein the carrierfrequency is between an adjacent pair of harmonics of 50, 60 or 400 Hz.14. An electrical circuit finder for detecting a conductive circuit, thefinder including: a transmitter for transmitting an electrical signal tothe conductive circuit and which includes: a modulator; and a reactiveload, the reactive load being coupled to the modulator; wherein theelectrical signal is generated across the load and is electricallycoupled to the conductive circuit; and a receiver which detects theelectrical signal in the conductive circuit.
 15. The electrical circuitfinder of claim 14, wherein the reactive load includes a first inductorand a second inductor, the first and second inductors are magneticallycoupled, and the first inductor is serially connected to the modulator.16. The electrical circuit finder of claim 15, wherein the finder is fordetecting a conductive circuit in an alternating current AC powersystem; and comprising a second modulator serially connected to thesecond inductor.
 17. The electrical circuit finder of claim 14, whereinthe reactive load includes a resonant LC circuit including a seriallyconnected inductor and capacitor.
 18. The electrical circuit finder ofclaim 17, wherein the modulator includes a switched-current sourceconnected in parallel to the capacitor.
 19. The electrical circuitfinder of claim 14, wherein the modulator operates at a carrierfrequency between approximately 120 Hz and approximately 3900 Hz. 20.The electrical circuit finder of claim 19, wherein the conductivecircuit is an AC power system; wherein the carrier frequency is betweenan adjacent pair of harmonics of the AC power system; and wherein thecarrier frequency is modulated by a sign function which represents asign of a voltage signal of the AC power system.
 21. An electricalcircuit finder for detecting a conductive circuit including: atransmitter for transmitting an electrical signal to the conductivecircuit; and a receiver which detects the electrical signal in theconductive circuit and includes a time-variant filter.
 22. Theelectrical circuit finder of claim 21, wherein the filter integrates foran integral number of cycles of 50, 60 or 400 Hz power system.
 23. Theelectrical circuit finder of claim 22, wherein the filter is an analogswitched-integrator.
 24. The electrical circuit finder of claim 22,wherein the filter is a digital filter.
 25. The electrical circuitfinder of claim 21, wherein the transmitter includes a modulator whichoperates at a carrier frequency between approximately 120 Hz andapproximately 3900 Hz.
 26. The electrical circuit finder of claim 22,wherein the conductive circuit carries alternating current from an ACpower system; wherein the carrier frequency is between an adjacent pairof harmonics of the AC power system; and wherein the carrier frequencyis modulated by a sign function representing a sign of a voltage signalof the AC power system.
 27. A method for finding an electrical circuitcomprising the acts of: connecting a transmitter to the electricalcircuit; inducing a phased-switched signal at a carrier frequency ontothe electrical circuit; sensing the electrical circuit with a receiverwhich detects the carrier frequency, wherein the sensing furtherincludes the acts of: generating a metric; saving the metric as areference; generating a next metric; comparing the reference to the nextmetric; and if the next metric is a set percentage of the reference,generating an alert indication to an operator; comparing the referenceto the next metric; and if the next metric is greater than thereference, setting the reference to the average of the next metric andthe reference; and repeating the acts of generating the next metric,comparing and generating the alert, and comparing and setting thereference.
 28. The method of claim 27, wherein the set percentage isapproximately between 70 to 95 percent.
 29. The method of claim 28,wherein the alert indication includes an optical indication.
 30. Themethod of claim 28, wherein the alert indication includes an audioindication.
 31. A method for finding an electrical circuit comprisingthe acts of: providing a carrier frequency between approximately 120 Hzand approximately 3900 Hz; modulating the carrier frequency; generatinga signal across a modulator and a load; coupling the signal to theelectrical circuit; and detecting the signal in the conductive circuit.